CS5127GDWR16 View Datasheet(PDF) - Cherry semiconductor

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CS5127GDWR16

Dual Output Nonsynchronous Buck Controller with Sync Function and Second Channel Enable

Cherry semiconductor 

Applications Information: continued

discharge time is typically less than 10% of the charge

time. External components CT and RT allow the switching

frequency to be set by the user in the range between 10kHz

and 500kHz. CT can be chosen first based on size and cost

constraints. For proper operation over temperature, the

value of RT should be chosen within the range from 20k½

to 40k½. Any type of one-eighth watt resistor will be ade-

quate. Larger values of RT will decrease the maximum

duty cycle slightly. This occurs because the sink current on

the CT lead has an exponential relationship to the charge

current. Higher charge currents will discharge the CT lead

capacitor more quickly than lower currents, and a shorter

discharge time will result in a higher maximum duty

cycle.

Once the oscillator frequency and a value of CT have been

selected, the necessary value of RT can be calculated as fol-

lows:

1.88

RT = (fOSC)(CT)

where fOSC is the oscillator frequency in hertz, CT is given

in farads, and the value of RT is given in ohms. ESR effects

are negligible since the charge and discharge currents are

fairly small, and any type of capacitor is adequate for CT.

Selecting the Compensation Capacitor

As previously noted, the error amplifier does not con-

tribute greatly to transient response, but it does influence

noise immunity. The fast feedback loop input is compared

against the COMP pin voltage. The DC bias to the VFFB pin

may be provided directly from the output voltage, or

through a resistor divider if output voltage is greater than

2.9V. The desired percentage value of DC accuracy trans-

lates directly to the VFFB pin, and the minimum COMP pin

capacitor value can be calculated:

(16mA)(TOSC)

CCOMP = (VFFBDC Bias Voltage)(tolerance)

If fOSC = 200kHz, VFFB DC bias voltage is 2.8V and toler-

ance is 0.1%, CCOMP = 28.6µF. This is the minimum value

of COMP pin capacitance that should be used. It is a good

practice to guard band the tolerance used in the calcula-

tion. Larger values of capacitance will improve noise

immunity, and a 100µF capacitor will work well in most

applications.

The type of capacitor is not critical, since the amplifier

output sink current of 16mA into a fairly large value or

wide range of ESR will typically result in a very small DC

output voltage error. The COMP pin capacitor also deter-

mines the length of the soft start interval.

Selecting the Input Bypass Capacitor

The input bypass capacitors minimize the ripple current in

the input supply, help to minimize EMI, and provide a

charge reservoir to improve transient response. The capac-

itor ripple current rating places the biggest constraint on

component selection. The input bypass capacitor network

should conduct all the ripple current. RMS ripple current

can be as large as half the load current, and can be calcu-

lated as:

IRIPPLE(RMS) = IOUT

VOUT(VIN - VOUT)

VIN2

Peak current requirement, load transients, ambient operat-

ing temperature and product reliability requirements all

play a role in choosing this component. Capacitor ESR and

the maximum load current step will determine the maxi-

mum transient variation of the supply voltage during

normal operation. The drop in the supply voltage due to

load transient response is given as:

ÆV = IRIPPLE(RMS) ´ ESR

The type of capacitor is also an important consideration.

Aluminum electrolytic capacitors are inexpensive, but they

typically have low ripple current ratings. Choosing larger

values of capacitance will increase the ripple current

rating, but physical size will increase as well. Size con-

straints may eliminate aluminum electrolytics fro

consideration. Aluminum electrolytics typically have

shorter operating life because the electrolyte evaporates

during operation. Tantalum electrolytic capacitors have

been associated with failure from inrush current, and man-

ufacturers of these components recommended derating the

capacitor voltage by a ratio 2:1 in surge applications. Some

manufacturers have product lines specifically tested to

withstand high inrush current. AVX TPS capacitors are

one such product. Ceramic capacitors perform well, but

they are also large and fairly expensive.

Startup

At startup, output switching does not occur until two

undervoltage lockouts release. The first lockout monitors

the VIN lead voltage. No internal IC activity occurs until

VIN lead voltage exceeds the VIN turn-on threshold. This

threshold is typically 8.4V. Once this condition is met, the

on-chip reference turns on. As the reference voltage begins

to rise, a second undervoltage lockout disables switching

until VREF lead voltage is about 3.5V. The GATE leads are

held in a low state until both lockouts are released.

As switching begins, the VFB lead voltage is lower than the

output voltage. This causes the error amplifier to source

current to the COMP lead capacitor. The COMP lead volt-

age will begin to rise. As the COMP lead voltage begins to

rise, it sets the threshold level at which the rising VFFB lead

voltage will trip the PWM comparator and terminate

switch conduction. This process results in a soft start inter-

val. The DC bias voltage on VFFB will determine the final

COMP voltage after startup, and the soft start time can be

approximately calculated as:

TSOFT START =

VFFB ´ CCOMP

ICOMP(SOURCE)

10

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