AD8314ACP-REEL_02 View Datasheet(PDF) - Analog Devices
Part Name
Description
MFG CO.
AD8314ACP-REEL_02 Datasheet PDF : 17 Pages
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AD8314
The relationship between the input level and the setpoint voltage
follows from the nominal transfer function of the device (VUP vs.
Input Amplitude, see TPC 1). For example, a voltage of 1 V on
VSET is demanding a power level of 0 dBm at RFIN. The corre-
sponding power level at the output of the power amplifier will be
greater than this amount due to the attenuation through the direc-
tional coupler.
When connected in a PA control loop, as shown in Figure 7, the
voltage VUP is not explicitly used, but is implicated in again setting
up the required averaging time, by choice of CF. However, now the
effective loop response time is a much more complicated function
of the PA’s gain-control characteristics, which are very nonlinear.
A complete solution requires specific knowledge of the power
amplifier.
The transient response of this control loop is determined by the
filter capacitor, CF. When this is large, the loop will be uncon-
ditionally stable (by virtue of the “dominant pole” generated
by this capacitor), but the response will be sluggish. The minimum
value ensuring stability should be used, requiring full attention
to the particulars of the power amplifier control function. Because
this is invariably nonlinear, the choice must be made for the
worst-case condition, which usually corresponds to the smallest
output from the PA, where the gain function is steepest. In practice,
an improvement in loop dynamics can often be achieved by adding
a response zero, formed by a resistor in series with CF.
Power-On and Enable Glitch
As already mentioned, the AD8314 can be put into a low power
mode by pulling the ENBL pin to ground. This reduces the quiescent
current from 4.5 mA to 20 µA. Alternatively, the supply can be
turned off completely to eliminate the quiescent current. TPCs 13
and 23 show the behavior of the V_DN output under these two
conditions (in TPC 23, ENBL is tied to VPOS). The glitch that
results in both cases can be reduced by loading the V_DN output.
In Figure 8b, the matching components are drawn as general
reactances. Depending on the frequency, the input impedance at
that frequency and the availability of standard value components,
either a capacitor or an inductor will be used. As in the previous
case, the input impedance at a particular frequency is plotted on
a Smith Chart and matching components are chosen (shunt
or series L, shunt or series C) to move the impedance to the
center of the chart. Table II gives standard component values
for some popular frequencies. Matching components for other
frequencies can be calculated using the input resistance and reac-
tance data over frequency which is given in TPC 9. Note that
the reactance is plotted as though it appears in parallel with the
input impedance (which it does because the reactance is primarily
due to input capacitance).
The impedance matching characteristics of a reactive matching
network provide voltage gain ahead of the AD8314; this increases
the device sensitivity (see Table II). The voltage gain is calculated
using the equation:
Voltage GaindB = 20 log10
R2
R1
where R2 is the input impedance of the AD8314 and R1 is the
source impedance to which the AD8314 is being matched. Note
that this gain will only be achieved for a perfect match. Component
tolerances and the use of standard values will tend to reduce
the gain.
50⍀ SOURCE
50⍀
AD8314
RFIN
RSHUNT
52.3⍀
CC
CIN RIN
VBIAS
Input Coupling Options
The internal 5 pF coupling capacitor of the AD8314, along with
the low frequency input impedance of 3 kΩ, gives a high-pass input
corner frequency of approximately 16 MHz. This sets the mini-
mum operating frequency. Figure 8 shows three options for
input coupling. A broadband resistive match can be implemented
by connecting a shunt resistor to ground at RFIN (Figure 8a).
This 52.3 Ω resistor (other values can also be used to select
different overall input impedances) combines with the input
impedance of the AD8314 (3 kΩʈ2 pF) to give a broadband
input impedance of 50 Ω. While the input resistance and capaci-
tance (CIN and RIN) will vary by approximately ± 20% from device
to device, the dominance of the external shunt resistor means
that the variation in the overall input impedance will be close
to the tolerance of the external resistor.
At frequencies above 2 GHz, the input impedance drops below
250 Ω (see TPC 9), so it is appropriate to use a larger value of
shunt resistor. This value is calculated by plotting the input
impedance (resistance and capacitance) on a Smith Chart and
choosing the best value of shunt resistor to bring the input imped-
ance closest to the center of the chart. At 2.5 GHz, a shunt
resistor of 165 Ω is recommended.
A reactive match can also be implemented as shown in Figure
8b. This is not recommended at low frequencies as device toler-
ances will dramatically vary the quality of the match because of
the large input resistance. For low frequencies, Option a or
Option c (see below) is recommended.
a. Broadband Resistive
50⍀ SOURCE
50⍀
AD8314
X1
RFIN
CC
X2
CIN RIN
VBIAS
b. Narrowband Reactive
STRIPLINE
RFIN
RATTN
AD8314
CC
CIN RIN
VBIAS
c. Series Attenuation
Figure 8. Input Coupling Options
Figure 8c shows a third method for coupling the input signal into
the AD8314, applicable in applications where the input signal
is larger than the input range of the log amp. A series resistor,
connected to the RF source, combines with the input impedance
of the AD8314 to resistively divide the input signal being applied
to the input. This has the advantage of very little power being
“tapped off” in RF power transmission applications.
REV. A
–11–
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