LTC1539 View Datasheet(PDF) - Linear Technology
Part Name
Description
MFG CO.
LTC1539 Datasheet PDF : 32 Pages
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1538-AUX/LTC1539 circuits. LTC1538-AUX/
LTC1539 VIN current, INTVCC current, I2R losses and
topside MOSFET transition losses.
1. The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET driver
and control currents. VIN current typically results in a
small (<< 1%) loss which increases with VIN.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC which is typically much larger than the
control circuit current. In continuous mode, IGATECHG =
f(QT + QB), where QT and QB are the gate charges of the
topside and bottom side MOSFETs. It is for this reason
that the large topside and synchronous MOSFETs are
turned off during low current operation in favor of the
small topside MOSFET and external Schottky diode,
allowing efficient, constant-frequency operation at low
output currents.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of Duty Cycle/
Efficiency. For example, in a 20V to 5V application,
10mA of INTVCC current results in approximately 3mA
of VIN current. This reduces the midcurrent loss from
10% or more (if the driver was powered directly from
VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense R. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside main
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then
the resistance of one MOSFET can simply be summed
with the resistances of L and RSENSE to obtain I2R
losses. For example, if each RDS(ON) = 0.05Ω, RL =
0.15Ω and RSENSE = 0.05Ω, then the total resistance is
0.25Ω. This results in losses ranging from 3% to 10%
as the output current increases from 0.5A to 2A. I2R
losses cause the efficiency to roll off at high output
currents.
4. Transition losses apply only to the topside MOSFET(s)
and only when operating at high input voltages (typically
20V or greater). Transition losses can be estimated from:
Transition Loss ≈ 2.5(VIN)1.85(IMAX)(CRSS)(f)
Other losses including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time,
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to (∆ILOAD)(ESR) where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal which
forces the regulator loop to adapt to the current change
and return VOUT to its steady-state value. During this
recovery time VOUT can be monitored for overshoot or
ringing which would indicate a stability problem. The ITH
external components shown in Figure 1 will prove ad-
equate compensation for most applications.
A second, more severe transient is caused by switching in
loads with large (> 1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25)(CLOAD).
Thus a 10µF capacitor would require a 250µs rise time,
limiting the charging current to about 200mA.
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